Dynamic igbt gate drive to reduce switching loss

ABSTRACT

A vehicle powertrain includes an IGBT that conducts current between a supply and load. The vehicle powertrain also includes a controller that applies voltage to a gate of the IGBT at a first level for a first duration that depends on a capacitance of the gate, and increases the voltage over a second duration based on a rate of change of the current falling below a threshold defined by a supply voltage for the load.

TECHNICAL FIELD

This application is generally related to control of a gate voltage of anIGBT in a hybrid-electric powertrain in which the gate voltage includesa step function maintained at a first level after which the gate voltageis increased over a period of time to a second voltage level.

BACKGROUND

Electrified vehicles including hybrid-electric vehicles (HEVs) andbattery electric vehicles (BEVs) rely on a traction battery to providepower to a traction motor for propulsion and a power invertertherebetween to convert direct current (DC) power to alternating current(AC) power. The typical AC traction motor is a 3-phase motor that may bepowered by 3 sinusoidal signals each driven with 120 degrees phaseseparation. The traction battery is configured to operate in aparticular voltage range. The terminal voltage of a typical tractionbattery is over 100 Volts DC and the traction battery is alternativelyreferred to as a high-voltage battery. However, improved performance ofelectric machines may be achieved by operating in a different voltagerange, typically at higher voltages than the traction battery. Manyelectrified vehicles include a DC-DC converter also referred to as avariable voltage converter (VVC) to convert the voltage of the tractionbattery to an operational voltage level of the electric machine. Theelectric machine may require a high voltage and high current. Due to thevoltage, current and switching requirements, an Insulated Gate Bipolarjunction Transistor (IGBT) is typically used to generate the signals inthe power inverter and the VVC.

SUMMARY

A vehicle powertrain includes an IGBT configured to conduct currentbetween a supply and load, and a controller configured to apply voltageto a gate of the IGBT at a first level for a first duration that dependson a capacitance of the gate, and to increase the voltage over a secondduration based on a rate of change of the current falling below athreshold defined by a supply voltage for the load.

A method of controlling an electric machine of a vehicle includes, by agate driver, applying a voltage at a first level onto a gate of an IGBTfor a predetermined time that is based on a capacitance of the gate,flowing, by the IGBT in response to the voltage, a current through aphase of the electric machine, and in response to a rate of change ofthe current through the phase exceeding a predetermined thresholddefined by a supply voltage of the electric machine, transitioning fromthe first level to a second level greater than the first level.

A vehicle includes an IGBT configured to selectively conduct currentbetween a supply and load, and a controller configured to apply avoltage to a gate of the IGBT at a first level for a duration derivedfrom a resistance of the gate, and after the duration expires, control arate of increase of the voltage based on a rate of change of the currentbeing less than a threshold corresponding to a supply voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a hybrid vehicle illustrating typical drivetrainand energy storage components with a power inverter therebetween.

FIG. 2 is a schematic of a vehicular variable voltage converter.

FIG. 3 is a schematic of a vehicular electric motor inverter.

FIG. 4 is a graphical representation of a gate voltage profile withrespect to time.

FIG. 5 is a flow diagram illustrating gate drive control for an IGBT.

FIG. 6A is a graphical representation of voltage profiles associatedwith a gate of an IGBT with respect to time.

FIG. 6B is a graphical representation of a current profile associatedwith a gate of an IGBT with respect to time.

FIG. 6C is a graphical representation of a collector to emitter voltageprofile associated with an IGBT with respect to time.

FIG. 6D is a graphical representation of a current profile associatedwith a collector of an IGBT with respect to time.

DETAILED DESCRIPTION

Embodiments of the present disclosure are described herein. It is to beunderstood, however, that the disclosed embodiments are merely examplesand other embodiments can take various and alternative forms. Thefigures are not necessarily to scale; some features could be exaggeratedor minimized to show details of particular components. Therefore,specific structural and functional details disclosed herein are not tobe interpreted as limiting, but merely as a representative basis forteaching one skilled in the art to variously employ the presentinvention. As those of ordinary skill in the art will understand,various features illustrated and described with reference to any one ofthe figures can be combined with features illustrated in one or moreother figures to produce embodiments that are not explicitly illustratedor described. The combinations of features illustrated providerepresentative embodiments for typical applications. Variouscombinations and modifications of the features consistent with theteachings of this disclosure, however, could be desired for particularapplications or implementations.

Insulated Gate Bipolar junction Transistors (IGBTs) and flyback orfreewheeling diodes are widely used in a variety of industrialapplications, such as electric motor drives and power inverters.Operation of an IGBT is controlled by a gate voltage supplied by a gatedriver. Conventional gate drivers are typically based on a voltage,greater than a threshold voltage, applied to an IGBT gate with a currentlimiting resistor, which consists of a switchable voltage source andgate resistor. A low gate resistance would lead to a fast switchingspeed and low switching loss, but also cause higher stresses on thesemiconductor devices, e.g. over-voltage stress. Therefore, the gateresistance is selected to seek a compromise between switching loss,switching delay, and stresses.

Some disadvantages associated with conventional gate drivers for IGBTturn-on include limited control of switching delay time, current slopeand voltage slope such that optimization switching losses is limited.Another disadvantage is that a gate resistance is typically selectedbased on worst case operating condition thus introducing excessiveswitching losses under normal operating conditions. For example, at ahigh dc bus voltage, a gate resistance is selected based on a change incurrent with respect to time (di/dt) in order to avoid excessive diodevoltage overshoot during diode fly-back of the load. However, at low dcbus voltage the use of the gate resistance selected to protect for highbus voltages introduces excessive switching losses as a switching speedis then reduced by the gate resistance even though diode over-voltage isbelow a threshold.

A smart gate driving strategy is critical to achieve optimal switchingperformance for the whole switching trajectory and over all theoperating ranges. Here, a proposed step-ramp voltage gate drivingstrategy with feedback of operating conditions (e.g., voltage, loadcurrent, temperature, etc.) for IGBT turn-on is presented. The gatevoltage initially corresponds to the IGBT being off. A controller thenreceives a signal to turn on the IGBT after which the controller appliesa voltage step function to the IGBT gate. The voltage step function isat a level above a threshold voltage and below a minimum gate voltagelevel at which the IGBT is operated in a saturation mode at which pointthe collector current of the IGBT is equal to a maximum load current.The voltage level is maintained at this level for a duration that isderived from device characteristics such as a gate capacitance or a gateresistance. At the end of the duration, the voltage is ramped to aIGBT-on gate voltage The gate voltage is ramped over a period of time,that is based on a derivative of the current being less than a thresholdcorresponding to a supply voltage.

The step function gate voltage is selected to reduce the turn-on delaytime, as well as increase switching speed and reduce switching loss. Theramped increase of the gate voltage slows down the switching speed toavoid the excessive voltage overshoot across freewheeling diode. Thetiming for each stage is adaptive to IGBT operating conditions, e.g.,switched voltage (Vce), to realize the optimal switching performanceover the whole operating ranges. The gate driver produces the highestgate voltage ramping rate based on the operating conditions, in order toachieve a minimum switching loss while keeping the diode voltageovershoot within safety limit

FIG. 1 depicts an electrified vehicle 112 that may be referred to as aplug-in hybrid-electric vehicle (PHEV). A plug-in hybrid-electricvehicle 112 may comprise one or more electric machines 114 mechanicallycoupled to a hybrid transmission 116. The electric machines 114 may becapable of operating as a motor or a generator. In addition, the hybridtransmission 116 is mechanically coupled to an engine 118. The hybridtransmission 116 is also mechanically coupled to a drive shaft 120 thatis mechanically coupled to the wheels 122. The electric machines 114 canprovide propulsion and deceleration capability when the engine 118 isturned on or off. The electric machines 114 may also act as generatorsand can provide fuel economy benefits by recovering energy that wouldnormally be lost as heat in a friction braking system. The electricmachines 114 may also reduce vehicle emissions by allowing the engine118 to operate at more efficient speeds and allowing the hybrid-electricvehicle 112 to be operated in electric mode with the engine 118 offunder certain conditions. An electrified vehicle 112 may also be abattery electric vehicle (BEV). In a BEV configuration, the engine 118may not be present. In other configurations, the electrified vehicle 112may be a full hybrid-electric vehicle (FHEV) without plug-in capability.

A traction battery or battery pack 124 stores energy that can be used bythe electric machines 114. The vehicle battery pack 124 may provide ahigh voltage direct current (DC) output. The traction battery 124 may beelectrically coupled to one or more power electronics modules 126. Oneor more contactors 142 may isolate the traction battery 124 from othercomponents when opened and connect the traction battery 124 to othercomponents when closed. The power electronics module 126 is alsoelectrically coupled to the electric machines 114 and provides theability to bi-directionally transfer energy between the traction battery124 and the electric machines 114. For example, a traction battery 124may provide a DC voltage while the electric machines 114 may operatewith a three-phase alternating current (AC) to function. The powerelectronics module 126 may convert the DC voltage to a three-phase ACcurrent to operate the electric machines 114. In a regenerative mode,the power electronics module 126 may convert the three-phase AC currentfrom the electric machines 114 acting as generators to the DC voltagecompatible with the traction battery 124.

The vehicle 112 may include a variable-voltage converter (VVC) 152electrically coupled between the traction battery 124 and the powerelectronics module 126. The VVC 152 may be a DC/DC boost converterconfigured to increase or boost the voltage provided by the tractionbattery 124. By increasing the voltage, current requirements may bedecreased leading to a reduction in wiring size for the powerelectronics module 126 and the electric machines 114. Further, theelectric machines 114 may be operated with better efficiency and lowerlosses.

In addition to providing energy for propulsion, the traction battery 124may provide energy for other vehicle electrical systems. The vehicle 112may include a DC/DC converter module 128 that converts the high voltageDC output of the traction battery 124 to a low voltage DC supply that iscompatible with low-voltage vehicle loads. An output of the DC/DCconverter module 128 may be electrically coupled to an auxiliary battery130 (e.g., 12V battery) for charging the auxiliary battery 130. Thelow-voltage systems may be electrically coupled to the auxiliary battery130. One or more electrical loads 146 may be coupled to the high-voltagebus. The electrical loads 146 may have an associated controller thatoperates and controls the electrical loads 146 when appropriate.Examples of electrical loads 146 may be a fan, an electric heatingelement and/or an air-conditioning compressor.

The electrified vehicle 112 may be configured to recharge the tractionbattery 124 from an external power source 136. The external power source136 may be a connection to an electrical outlet. The external powersource 136 may be electrically coupled to a charger or electric vehiclesupply equipment (EVSE) 138. The external power source 136 may be anelectrical power distribution network or grid as provided by an electricutility company. The EVSE 138 may provide circuitry and controls toregulate and manage the transfer of energy between the power source 136and the vehicle 112. The external power source 136 may provide DC or ACelectric power to the EVSE 138. The EVSE 138 may have a charge connector140 for plugging into a charge port 134 of the vehicle 112. The chargeport 134 may be any type of port configured to transfer power from theEVSE 138 to the vehicle 112. The charge port 134 may be electricallycoupled to a charger or on-board power conversion module 132. The powerconversion module 132 may condition the power supplied from the EVSE 138to provide the proper voltage and current levels to the traction battery124. The power conversion module 132 may interface with the EVSE 138 tocoordinate the delivery of power to the vehicle 112. The EVSE connector140 may have pins that mate with corresponding recesses of the chargeport 134. Alternatively, various components described as beingelectrically coupled or connected may transfer power using a wirelessinductive coupling.

One or more wheel brakes 144 may be provided for decelerating thevehicle 112 and preventing motion of the vehicle 112. The wheel brakes144 may be hydraulically actuated, electrically actuated, or somecombination thereof. The wheel brakes 144 may be a part of a brakesystem 150. The brake system 150 may include other components to operatethe wheel brakes 144. For simplicity, the figure depicts a singleconnection between the brake system 150 and one of the wheel brakes 144.A connection between the brake system 150 and the other wheel brakes 144is implied. The brake system 150 may include a controller to monitor andcoordinate the brake system 150. The brake system 150 may monitor thebrake components and control the wheel brakes 144 for vehicledeceleration. The brake system 150 may respond to driver commands andmay also operate autonomously to implement features such as stabilitycontrol. The controller of the brake system 150 may implement a methodof applying a requested brake force when requested by another controlleror sub-function.

Electronic modules in the vehicle 112 may communicate via one or morevehicle networks. The vehicle network may include a plurality ofchannels for communication. One channel of the vehicle network may be aserial bus such as a Controller Area Network (CAN). One of the channelsof the vehicle network may include an Ethernet network defined byInstitute of Electrical and Electronics Engineers (IEEE) 802 family ofstandards. Additional channels of the vehicle network may includediscrete connections between modules and may include power signals fromthe auxiliary battery 130. Different signals may be transferred overdifferent channels of the vehicle network. For example, video signalsmay be transferred over a high-speed channel (e.g., Ethernet) whilecontrol signals may be transferred over CAN or discrete signals. Thevehicle network may include any hardware and software components thataid in transferring signals and data between modules. The vehiclenetwork is not shown in FIG. 1 but it may be implied that the vehiclenetwork may connect to any electronic module that is present in thevehicle 112. A vehicle system controller (VSC) 148 may be present tocoordinate the operation of the various components.

FIG. 2 depicts a diagram of a VVC 152 that is configured as a boostconverter. The VVC 152 may include input terminals that may be coupledto terminals of the traction battery 124 through the contactors 142. TheVVC 152 may include output terminals coupled to terminals of the powerelectronics module 126. The VVC 152 may be operated to cause a voltageat the output terminals to be greater than a voltage at the inputterminals. The vehicle 112 may include a VVC controller 200 thatmonitors and controls electrical parameters (e.g., voltage and current)at various locations within the VVC 152. In some configurations, the VVCcontroller 200 may be included as part of the VVC 152. The VVCcontroller 200 may determine an output voltage reference, V*_(dc). TheVVC controller 200 may determine, based on the electrical parameters andthe voltage reference, V*_(dc), a control signal sufficient to cause theVVC 152 to achieve the desired output voltage. In some configurations,the control signal may be implemented as a pulse-width modulated (PWM)signal in which a duty cycle of the PWM signal is varied. The controlsignal may be operated at a predetermined switching frequency. The VVCcontroller 200 may command the VVC 152 to provide the desired outputvoltage using the control signal. The particular control signal at whichthe VVC 152 is operated may be directly related to the amount of voltageboost to be provided by the VVC 152.

The output voltage of the VVC 152 may be controlled to achieve a desiredreference voltage. In some configurations, the VVC 152 may be a boostconverter. In a boost converter configuration in which the VVCcontroller 200 controls the duty cycle, the ideal relationship betweenthe input voltage V_(in) and the output voltage V_(out) and the dutycycle D may be illustrated using the following equation:

$\begin{matrix}{V_{out} = \frac{V_{in}}{\left( {1 - D} \right)}} & (1)\end{matrix}$

The desired duty cycle, D, may be determined by measuring the inputvoltage (e.g., traction battery voltage) and setting the output voltageto the reference voltage. The VVC 152 may be a buck converter thatreduces the voltage from input to output. In a buck configuration, adifferent expression relating the input and output voltage to the dutycycle may be derived. In some configurations, the VVC 152 may be abuck-boost converter that may increase or decrease the input voltage.The control strategy described herein is not limited to a particularvariable voltage converter topology.

With reference to FIG. 2, the VVC 152 may boost or “step up” the voltagepotential of the electrical power provided by the traction battery 124.The traction battery 124 may provide high voltage (HV) DC power. In someconfigurations, the traction battery 124 may provide a voltage between150 and 400 Volts. The contactor 142 may be electrically coupled inseries between the traction battery 124 and the VVC 152. When thecontactor 142 is closed, the HV DC power may be transferred from thetraction battery 124 to the VVC 152. An input capacitor 202 may beelectrically coupled in parallel to the traction battery 124. The inputcapacitor 202 may stabilize the bus voltage and reduce any voltage andcurrent ripple. The VVC 152 may receive the HV DC power and boost or“step up” the voltage potential of the input voltage according to theduty cycle.

An output capacitor 204 may be electrically coupled between the outputterminals of the VVC 152. The output capacitor 204 may stabilize the busvoltage and reduce voltage and current ripple at the output of the VVC152.

Further with reference to FIG. 2, the VVC 152 may include a firstswitching device 206 and a second switching device 208 for boosting aninput voltage to provide the boosted output voltage. The switchingdevices 206, 208 may be configured to selectively flow a current to anelectrical load (e.g., power electronics module 126 and electricmachines 114). Each switching device 206, 208 may be individuallycontrolled by a gate drive circuit (not shown) of the VVC controller 200and may include any type of controllable switch (e.g., an insulated gatebipolar transistor (IGBT) or field-effect transistor (FET)). The gatedrive circuit may provide electrical signals to each of the switchingdevices 206, 208 that are based on the control signal (e.g., duty cycleof PWM control signal). A diode may be coupled across each of theswitching devices 206, 208. The switching devices 206, 208 may each havean associated switching loss. The switching losses are those powerlosses that occur during state changes of the switching device (e.g.,on/off and off/on transitions). The switching losses may be quantifiedby the current flowing through and the voltage across the switchingdevice 206, 208 during the transition. The switching devices may alsohave associated conduction losses that occur when the device is switchedon.

The vehicle system may include sensors for measuring electricalparameters of the VVC 152. A first voltage sensor 210 may be configuredto measure the input voltage, (e.g., voltage of the battery 124), andprovide a corresponding input signal (V_(bat)) to the VVC controller200. In one or more embodiments, the first voltage sensor 210 maymeasure the voltage across the input capacitor 202, which corresponds tothe battery voltage. A second voltage sensor 212 may measure the outputvoltage of the VVC 152 and provide a corresponding input signal (V_(dc))to the VVC controller 200. In one or more embodiments, the secondvoltage sensor 212 may measure the voltage across the output capacitor204, which corresponds to the DC bus voltage. The first voltage sensor210 and the second voltage sensor 212 may include circuitry to scale thevoltages to a level appropriate for the VVC controller 200. The VVCcontroller 200 may include circuitry to filter and digitize the signalsfrom the first voltage sensor 210 and the second voltage sensor 212.

An input inductor 214 may be electrically coupled in series between thetraction battery 124 and the switching devices 206, 208. The inputinductor 214 may alternate between storing and releasing energy in theVVC 152 to enable the providing of the variable voltages and currents asVVC 152 output, and the achieving of the desired voltage boost. Acurrent sensor 216 may measure the input current through the inputinductor 214 and provide a corresponding current signal (I_(L)) to theVVC controller 200. The input current through the input inductor 214 maybe a result of the voltage difference between the input and the outputvoltage of the VVC 152, the conducting time of the switching devices206, 208, and the inductance L of the input inductor 214. The VVCcontroller 200 may include circuitry to scale, filter, and digitize thesignal from the current sensor 216. In another embodiment, a bypassdiode 218 may be coupled between the input of the VVC and the output ofthe VVC such that the output of the VVC (e.g., inverter input voltage)is clamped to the input voltage of the VVC (e.g., the traction batteryvoltage).

The VVC controller 200 may be programmed to control the output voltageof the VVC 152. The VVC controller 200 may receive input from the VVC152 and other controllers via the vehicle network, and determine thecontrol signals. The VVC controller 200 may monitor the input signals(V_(bat), V_(dc), I_(L),V*_(dc)) to determine the control signals. Forexample, the VVC controller 200 may provide control signals to the gatedrive circuit that correspond to a duty cycle command. The gate drivecircuit may then control each switching device 206, 208 based on theduty cycle command.

The control signals to the VVC 152 may be configured to drive theswitching devices 206, 208 at a particular switching frequency. Withineach cycle of the switching frequency, the switching devices 206, 208may be operated at the specified duty cycle. The duty cycle defines theamount of time that the switching devices 206, 208 are in an on-stateand an off-state. For example, a duty cycle of 100% may operate theswitching devices 206, 208 in a continuous on-state with no turn off. Aduty cycle of 0% may operate the switching devices 206, 208 in acontinuous off-state with no turn on. A duty cycle of 50% may operatethe switching devices 206, 208 in an on-state for half of the cycle andin an off-state for half of the cycle. The control signals for the twoswitches 206, 208 may be complementary. That is, the control signal sentto one of the switching devices (e.g., 206) may be an inverted versionof the control signal sent to the other switching device (e.g., 208).

The current that is controlled by the switching devices 206, 208 mayinclude a ripple component that has a magnitude that varies with amagnitude of the current, and the duty cycle and switching frequency ofthe switching devices 206, 208. Relative to the input current, the worstcase ripple current magnitude occurs during relatively high inputcurrent conditions. When the duty cycle is fixed, an increase in theinductor current causes an increase in magnitude of the ripple currentas illustrated in FIG. 4. The magnitude of the ripple current is alsorelated to the duty cycle. The highest magnitude ripple current occurswhen the duty cycle equals 50%. The general relationship between theinductor ripple current magnitude and the duty cycle may be as shown inFIG. 5. Based on these facts, it may be beneficial to implement measuresto reduce the ripple current magnitude under high current and mid-rangeduty cycle conditions.

When designing the VVC 152, the switching frequency and the inductancevalue of the inductor 214 may be selected to satisfy a maximum allowableripple current magnitude. The ripple component may be a periodicvariation that appears on a DC signal. The ripple component may bedefined by a ripple component magnitude and a ripple componentfrequency. The ripple component may have harmonics that are in anaudible frequency range that may add to the noise signature of thevehicle. Further, the ripple component may cause difficulties withaccurately controlling devices fed by the source. During switchingtransients, the switching devices 206, 208 may turn off at the maximuminductor current (DC current plus ripple current) which may cause largevoltage spike across the switching devices 206, 208. Because of size andcost constraints, the inductance value may be selected based on theconducted current. In general, as current increases the inductance maydecrease due to saturation.

The switching frequency may be selected to limit a magnitude of theripple current component under worst case scenarios (e.g., highest inputcurrent and/or duty cycle close to 50% conditions). The switchingfrequency of the switching devices 206, 208 may be selected to be afrequency (e.g., 10kHz) that is greater than a switching frequency ofthe motor/generator inverter (e.g., 5kHz) that is coupled to an outputof the VVC 152. In some applications, the switching frequency of the VVC152 may be selected to be a predetermined fixed frequency. Thepredetermined fixed frequency is generally selected to satisfy noise andripple current specifications. However, the choice of the predeterminedfixed frequency may not provide best performance over all operatingranges of the VVC 152. The predetermined fixed frequency may providebest results at a particular set of operating conditions, but may be acompromise at other operating conditions.

Increasing the switching frequency may decrease the ripple currentmagnitude and lower voltage stress across the switching devices 206,208, but may lead to higher switching losses. While the switchingfrequency may be selected for worst case ripple conditions, the VVC 152may only operate under the worst case ripple conditions for a smallpercentage of the total operating time. This may lead to unnecessarilyhigh switching losses that may lower fuel economy. In addition, thefixed switching frequency may concentrate the noise spectrum in a verynarrow range. The increased noise density in this narrow range mayresult in noticeable noise, vibration, and harshness (NVH) issues.

The VVC controller 200 may be programmed to vary the switching frequencyof the switching devices 206, 208 based on the duty cycle and the inputcurrent. The variation in switching frequency may improve fuel economyby reducing switching losses and reduce NVH issues while maintainingripple current targets under worst case operating conditions.

During relatively high current conditions, the switching devices 206,208 may experience increased voltage stress. At a maximum operatingcurrent of the VVC 152, it may be desired to select a relatively highswitching frequency that reduces the ripple component magnitude with areasonable level of switching losses. The switching frequency may beselected based on the input current magnitude such that as the inputcurrent magnitude increases, the switching frequency increases. Theswitching frequency may be increased up to a predetermined maximumswitching frequency. The predetermined maximum switching frequency maybe a level that provides a compromise between lower ripple componentmagnitudes and higher switching losses. The switching frequency may bechanged in discrete steps or continuously over the operating currentrange.

The VVC controller 200 may be programmed to reduce the switchingfrequency in response to the current input being less than apredetermined maximum current. The predetermined maximum current may bea maximum operating current of the VVC 152. The change in the switchingfrequency may be based on the magnitude of the current input to theswitching devices 206, 208. When the current is greater than thepredetermined maximum current, the switching frequency may be set to apredetermined maximum switching frequency. As the current decreases, themagnitude of the ripple component decreases. By operating at lowerswitching frequencies as the current decreases, switching losses arereduced. The switching frequency may be varied based on the power inputto the switching devices. As the input power is a function of the inputcurrent and the battery voltage, the input power and input current maybe used in a similar manner.

Since the ripple current is also affected by the duty cycle, theswitching frequency may be varied based on the duty cycle. The dutycycle may be determined based on a ratio of the input voltage to theoutput voltage. As such, the switching frequency may also be variedbased on the ratio between the input voltage and the output voltage.When the duty cycle is near 50%, the predicted ripple current magnitudeis a maximum value and the switching frequency may be set to thepredetermined maximum frequency. The predetermined maximum frequency maybe a maximum switching frequency value that is selected to minimize theripple current magnitude. The switching frequency may be changed indiscrete steps or continuously over the duty cycle range.

The VVC controller 200 may be programmed to reduce the switchingfrequency from the predetermined maximum frequency in response to amagnitude of a difference between the duty cycle and the duty cyclevalue (e.g, 50%) at which the predicted ripple component magnitude is amaximum. When the magnitude of the difference is less than a threshold,the switching frequency may be set to the predetermined frequency. Whenthe magnitude of the difference decreases, the switching frequency maybe increased toward the predetermined maximum frequency to reduce theripple component magnitude. When the magnitude of the difference is lessthan a threshold, the switching frequency may be set to thepredetermined maximum frequency.

The switching frequency may be limited to be between the predeterminedmaximum frequency and a predetermined minimum frequency. Thepredetermined minimum frequency may be a frequency level that is greaterthan a predetermined switching frequency of the power electronic module126 that is coupled to an output of the voltage converter 152.

With reference to FIG. 3, a system 300 is provided for controlling apower electronics module (PEM) 126. The PEM 126 of FIG. 3 is shown toinclude a plurality of switches 302 (e.g., IGBTs) configured tocollectively operate as an inverter with first, second, and third phaselegs 316, 318, 320. While the inverter is shown as a three-phaseconverter, the inverter may include additional phase legs. For example,the inverter may be a four-phase converter, a five-phase converter, asix-phase converter, etc. In addition, the PEM 126 may include multipleconverters with each inverter in the PEM 126 including three or morephase legs. For example, the system 300 may control two or moreinverters in the PEM 126. The PEM 126 may further include a DC to DCconverter having high power switches (e.g., IGBTs) to convert a powerelectronics module input voltage to a power electronics module outputvoltage via boost, buck or a combination thereof.

As shown in FIG. 3, the inverter may be a DC-to-AC converter. Inoperation, the DC-to-AC converter receives DC power from a DC power link306 through a DC bus 304 and converts the DC power to AC power. The ACpower is transmitted via the phase currents ia, ib, and is to drive anAC machine also referred to as an electric machine 114, such as athree-phase permanent-magnet synchronous motor (PMSM) as depicted inFIG. 3. In such an example, the DC power link 306 may include a DCstorage battery to provide DC power to the DC bus 304. In anotherexample, the inverter may operate as an AC-to-DC converter that convertsAC power from the AC machine 114 (e.g., generator) to DC power, whichthe DC bus 304 can provide to the DC power link 306. Furthermore, thesystem 300 may control the PEM 126 in other power electronic topologies.

With continuing reference to FIG. 3, each of the phase legs 316, 318,320 in the inverter includes power switches 302, which may beimplemented by various types of controllable switches. In oneembodiment, each power switch 302 may include a diode and a transistor,(e.g., an IGBT). The diodes of FIG. 3 are labeled D_(a1), D_(a2),D_(b1), D_(b2), D_(c1), and D_(c2) while the IGBTs of FIG. 3 arerespectively labeled S_(a1), S_(a2), S_(b1), S_(b2), S_(c1), and S_(c2).The power switches S_(a1,) S_(a2), D_(a1), and D_(a2) are part of phaseleg A of the three-phase converter, which is labeled as the first phaseleg a 316 in FIG. 3. Similarly, the power switches S_(b1), S_(b2),D_(b1), and D_(b2) are part of phase leg B 318 and the power switchesS_(c1), S_(c2), D_(c1), and D_(c2) are part of phase leg C 320 of thethree-phase converter. The inverter may include any number of the powerswitches 302 or circuit elements depending on the particularconfiguration of the inverter. The diodes (D_(xx)) are connected inparallel with the IGBTs (S_(xx)) however, as the polarities are reversedfor proper operation, this configuration is often referred to as beingconnected anti-parallel. A diode in this anti-parallel configuration isalso called a freewheeling diode.

As illustrated in FIG. 3, current sensors CS_(a), CS_(b), and CS_(c) areprovided to sense current flow in the respective phase legs 316, 318,320. FIG. 3 shows the current sensors CS_(a), CS_(b), and CS_(c)separate from the PEM 126. However, current sensors CS_(a), CS_(b), andCS_(c) may be integrated as part of the PEM 126 depending on itsconfiguration. Current sensors CS_(a), CS_(b), and CS_(c) of FIG. 3 areinstalled in series with each of phase legs A, B and C (i.e., phase legs316, 318, 320 in FIG. 3) and provide the respective feedback signalsi_(as), i_(bs), and i_(cs)(also illustrated in FIG. 3) for the system300. The feedback signals i_(as), i_(bs), and i_(cs) may be raw currentsignals processed by logic device (LD) 310 or may be embedded or encodedwith data or information about the current flow through the respectivephase legs 316, 318, 320. Also, the power switches 302 (e.g., IGBTs) mayinclude current sensing capability. The current sensing capability mayinclude being configured with a current mirror output, which may providedata/signals representative of i_(as), i_(bs), and i_(cs). Thedata/signals may indicate a direction of current flow, a magnitude ofcurrent flow, or both the direction and magnitude of current flowthrough the respective phase legs A, B, and C.

Referring again to FIG. 3, the system 300 includes a logic device (LD)or controller 310. The controller or LD 310 can be implemented byvarious types or combinations of electronic devices and/ormicroprocessor-based computers or controllers. To implement a method ofcontrolling the PEM 126, the controller 310 may execute a computerprogram or algorithm embedded or encoded with the method and stored involatile and/or persistent memory 312. Alternatively, logic may beencoded in discrete logic, a microprocessor, a microcontroller, or alogic or gate array stored on one or more integrated circuit chips. Asshown in the embodiment of FIG. 3, the controller 310 receives andprocesses the feedback signals i_(as), i_(bs), and i_(cs) to control thephase currents i_(a), i_(b), and i_(c) such that the phase currentsi_(a), i_(b), and i_(c) flow through the phase legs 316, 318, 320 andinto the respective windings of the electric machine 114 according tovarious current or voltage patterns. For example, current patterns caninclude patterns of phase currents i_(a), i_(b), and i_(c) flowing intoand away from the DC-bus 304 or a DC-bus capacitor 308. The DC-buscapacitor 308 of FIG. 3 is shown separate from the PEM 126. However, theDC-bus capacitor 308 may be integrated as part of the PEM 126.

As shown in FIG. 3, a storage medium 312 (hereinafter “memory”), such ascomputer-readable memory may store the computer program or algorithmembedded or encoded with the method. In addition, the memory 312 maystore data or information about the various operating conditions orcomponents in the PEM 126. For example, the memory 312 may store data orinformation about current flow through the respective phase legs 316,318, 320. The memory 312 can be part of the controller 310 as shown inFIG. 3. However, the memory 312 may be positioned in any suitablelocation accessible by the controller 310.

As illustrated in FIG. 3, the controller 310 transmits at least onecontrol signal 236 to the power converter system 212. The powerconverter system 212 receives the control signal 322 to control theswitching configuration of the inverter and therefore the current flowthrough the respective phase legs 316, 318, and 320. The switchingconfiguration is a set of switching states of the power switches 302 inthe inverter. In general, the switching configuration of the inverterdetermines how the inverter converts power between the DC power link 306and the electric machine 114.

To control the switching configuration of the inverter, the inverterchanges the switching state of each power switch 302 in the inverter toeither an ON state or an OFF state based on the control signal 322. Inthe illustrated embodiment, to switch the power switch 302 to either ONor OFF states, the controller/LD 310 provides the gate voltage (Vg) toeach power switch 302 and therefore drives the switching state of eachpower switch 302. Gate voltages Vg_(a1), Vg_(a2), Vg_(b1), Vg_(b2),Vg_(c1), and Vg_(c2) (shown in FIG. 3) control the switching state andcharacteristics of the respective power switches 302. While the inverteris shown as a voltage-driven device in FIG. 3, the inverter may be acurrent-driven device or controlled by other strategies that switch thepower switch 302 between ON and OFF states. The controller 310 maychange the gate drive for each IGBT based on the rotational speed of theelectric machine 114, the mirror current, or a temperature of the IGBTswitch. The change in gate drive may be selected from a plurality ofgate drive currents in which the change gate drive current isproportional to a change in IGBT switching speed.

As also shown in FIG. 3, each phase leg 316, 318, and 320 includes twoswitches 302. However, only one switch in each of the legs 316, 318, 320can be in the ON state without shorting the DC power link 306. Thus, ineach phase leg, the switching state of the lower switch is typicallyopposite the switching state of the corresponding upper switch.Consequently, a HIGH state of a phase leg refers to the upper switch inthe leg in the ON state with the lower switch in the OFF state.Likewise, a LOW state of the phase leg refers to the upper switch in theleg in the OFF state with the lower switch in the ON state. As a result,IGBTs with current mirror capability may be on all IGBTs, a subset ofIGBTs (e.g., S_(a1), S_(b1), S_(c1)) or a single IGBT.

Two situations can occur during an active state of the three-phaseconverter example illustrated in FIG. 2: (1) two phase legs are in theHIGH state while the third phase leg is in the LOW state, or (2) onephase leg is in the HIGH state while the other two phase legs are in theLOW state. Thus, one phase leg in the three-phase converter, which maybe defined as the “reference” phase for a specific active state of theinverter, is in a state opposite to the other two phase legs, or“non-reference” phases, that have the same state. Consequently, thenon-reference phases are either both in the HIGH state or both in theLOW state during an active state of the inverter.

FIG. 4 is an example graphical representation 400 of a profile 406 of agate voltage 402 with respect to time 404. Here, the profile 406 beginsat a gate voltage level in which the IGBT is off (V_(g) _(_) _(off)).The gate voltage at which the IGBT is off (V_(g) _(_) _(off)) is a gatevoltage below a voltage threshold at which the IGBT conducts a currentflow (V_(ge(th))). V_(gc(th)) is a gate voltage that initiates a flow ofcollector current greater than a leakage current. V_(ge(th)) istemperature dependent typically drops 10 to 20 mV per degree Celsius. Attime 410, a voltage step function is applied to the gate of the IGBT inwhich the step voltage level (V_(g) _(_) _(step1)) 412 is a gate voltagelevel above V_(ge(th)) and below a minimum gate voltage level at whichthe IGBT is operated in a saturation mode at which point the collectorcurrent of the IGBT is equal to a maximum load current. The voltagelevel is maintained at the step voltage level (V_(g) _(_) _(step1)) 412for a duration that is derived from a capacitance of the gate, and agate resistance. For example, the maximum time or duration that the gatevoltage is maintained at V_(g) _(_) _(step1) may follow equation 2.

$\begin{matrix}{t_{1{\_ \max}} = {{{- R_{g}} \cdot C_{ies} \cdot \ln}\frac{V_{g\_ {on}} - v_{{ge}{({th})}}}{V_{g\_ {on}} - V_{g\_ {off}}}}} & (2)\end{matrix}$

In which C_(ies) is the input capacitance of the IGBT, namely thecapacitance between the gate and emitter and the capacitance between thegate and collector, R_(g) is the gate resistance, and t_(1 max) is theduration V_(g) _(_) _(step1) is maintained. The duration (t₁ _(max) orT₁) V_(g) _(_) _(step1) is maintained is a ramp start time 414 minus thevoltage step function time 410.

At ramp start time 414, the voltage level applied to the gate of theIGBT is increased to a gate on voltage (V_(g) _(on) ) 418. V_(g) _(_)_(on) 418 is the gate voltage in which the IGBT is operated inconduction mode, is a gate voltage high enough to activate the IGBT inthe saturation region, and is a gate voltage that does not exceed a gatebreakdown voltage. Typically, V_(g) _(_) _(on) 418 is approximately 15V.The gate voltage ramps to V_(g) _(_) _(on) over a period of time, theperiod (T₂) is the ramp end time 416 minus the ramp start time 414. T₂is based on a derivative of the current being less than a thresholdcorresponding to a supply voltage. Generally, the IGBT switching loss isa function of the period T₂, in which the switching loss increases asthe period T₂ increases. Also, the current rising slope or rate ofchange is a function of T₂, as T₂ increases, the rate of change of thecurrent decreases. Further, the threshold is a function of the supplyvoltage in which the rate of change is dependent upon the supplyvoltage, which is also referred to as a DC bus voltage or DC linkvoltage. The rate of change may include an instantaneous rate of changewhich can be mathematically expressed as a derivative. Here, theinstantaneous rate of change is limited by the ability of a controllerto sample the signals to determine the derivative (e.g., actually takenover a short time interval small sample, such as a 10 ns sample andconversion time). For example, a supply voltage of 400V/300A may limitthe rate of change to 5 A/nS thus limit the duration T₂ to 0.30 uS.However, for lower supply voltages such as 300V/300 A, the rate ofchange may be limited to 6.6 A/nS and limit the duration T₂ to 0.10 uS.Based on this, the duration T₂ may be determined for different dc busvoltages Vdc or as a function of varying dc bus voltages, currents thatare flowing through the IGBT, and IGBT temperatures. In one embodiment,a look-up table is stored in non-volatile memory accessible by thecontroller. The controller may then select a reference value of T₂ basedon the sensed signals of voltage, current, temperature. The controllermay then send a command indicative of the length of T₂ to gate driver.

FIG. 5 is a flow diagram 500 illustrating gate drive control for anIGBT. In operation 502, a controller receives a turn on signalindicative of a command to turn on an IGBT device. The flow diagram 500may be implemented by a variety of methods including a table look-up, anopen-loop control system, a closed-loop control system, an analogcontrol system, a digital control system, an adaptive control system, afuzzy logic control system, and a neural network system.

In operation 502, a controller receives a signal to turn on an IGBTdevice after reception of the signal, the controller proceeds tooperation 504. Prior to turning on the IGBT, the gate voltage for theIGBT is off (e.g., V_(g) _(_) _(off)) that is a gate voltage is at alevel below V_(ge(th)).

In operation 504, the controller determines an initial gate voltagelevel and steps the gate voltage level to the predetermined level (e.g.,V_(g) _(_) _(step 1)). Generally, a turn-on switching loss is inverselyproportional to the gate current level and the gate voltage level. Avoltage applied to the gate at a gate voltage conduction thresholdV_(ge(th)) will create a conductive channel between the collector andemitter of the IGBT, however, the IGBT may be operated in a linearregion and therefore have a higher switching loss. Also, the initialgate voltage may also be limited by gate driver capabilities, as thecost associated with the gate driver is typically directly proportionalwith the gate driver capabilities. For example, a gate driver withhigher driving capabilities is typically more expensive than a gatedriver with lower driving capabilities. Here, a voltage step function isapplied to the gate of the IGBT at a step voltage level (e.g., V_(g)_(_) _(step1)). The step voltage level is a gate voltage level aboveV_(ge(th)) and below a minimum gate voltage level at which the IGBT isoperated to conduct a maximum load current to a load in a saturationmode. The step voltage level may be predetermined and stored in alook-up table so that the level may be easily determined based onoperating characteristics such as an IGBT temperature or calculated onthe fly using a method described above. After the voltage step isapplied, the controller proceeds to operation 506.

In operation 506, the controller maintains the voltage level at the stepvoltage level (e.g., V_(g) _(_) _(step1)) for a duration that is derivedfrom IGBT characteristics including a capacitance associated with thegate (e.g., a capacitance between the gate and emitter C_(ge), acapacitance between the gate and collector C_(gc), a capacitance betweenthe collector and emitter C_(ce), or a combination thereof), a gateresistance, a gate charge associated with operation of the IGBT (e.g.,total gate charge Q_(g)), an IGBT temperature, or switchingcharacteristics (e.g., a turn-on delay time, a turn-off delay time, andswitching losses). The duration may be predetermined and stored in alook-up table so that the level may be easily determined based on theIGBT operating characteristics or calculated on the fly using a methoddescribed above. Once the duration expires, (i.e., the initial gatevoltage has been maintained for the required duration), the controllerproceeds to operation 508.

In operation 508, the controller increases the voltage at a rate ofincrease. The rate of increase may be based on many factors includingdevice characteristics, operating conditions, and load characteristics.The device characteristics include a gate resistance, a gatecapacitance, a threshold voltage, a max collector current, a diodeforward current, and other IGBT characteristics. The operatingconditions include a temperature, a switching speed, a PWM duty cycle, asupply voltage, and a vehicle speed. The load characteristics include aninductance, a resistance, a max current, rotational speed, potentialenergy, kinetic energy, and other electrical or electro-mechanicalcharacteristics. The rate of increase is determined by the differencebetween the first step voltage (e.g., V_(g) _(_) _(step1)') and theIGBT-on voltage (e.g., V_(g) _(_) _(on)) and the duration over which thechange in voltage occurs. The duration is based on a derivative of thecurrent flowing between the collector of the IGBT and the emitter of theIGBT being less than a threshold corresponding to a supply voltage.

Generally, the duration (T₂) is associated with a ramp rate that isselected to avoid the excessive voltage overshoot across freewheelingdiode across all operating ranges. A worst case operating condition(e.g., maximum dc bus voltage) is selected to determine the duration. Inpractice, once a baseline is determined the duration may be increasedgradually until a voltage spike across the freewheeling diode exceeds alimit. This data may be used to form a look-up table based on supplyvoltage and temperature. Therefore, the duration may be determined basedon a DC bus voltage for an inverter, or a battery voltage for a DC-DCconverter.

In operation 510, the controller compares the voltage of the gate of theIGBT with a IGBT on gate voltage. If the voltage applied to the gate isless than a turned-on gate voltage level, the controller loops back tooperation 508. If the voltage applied to the gate equals an IGBT-on gatevoltage level (e.g., V_(g) _(_) _(on), the controller proceeds tooperation 512. In operation 512, the controller maintains the gatevoltage at V_(g) _(_) _(on) until another signal is received. V_(g) _(_)_(on) is the gate voltage in which the IGBT conducts a current betweenthe collector and emitter in a saturation mode and that the gate voltagethat does not exceed a gate breakdown voltage. Typically, V_(g) _(_)_(on) is approximately 15V.

FIGS. 6A-6D are graphical representations of IGBT operatingcharacteristics associated with a gate drive circuit with respect totime. The gate driving strategy is adaptive to different operatingconditions based on operational feedback. The switching behavior atdifferent dc bus voltages are shown in FIGS. 6A-6D. Switching the IGBTat lower dc bus voltages allows a higher collector current and a higherrate of change of collector current with respect to time (di/dt)considering the diode has more safety margin. Thus, the gate voltage(V_(g)) increases at a higher slope, which produces a larger gatecurrent (Ig) during current rising period to speed up the switching andreduce the loss. In this way, the step-ramp voltage source gate drivingstrategy optimizes switching behavior over all the operating ranges.Generally, a higher IGBT di/dt will lead to higher diode voltage spikes,which may subject diode to over-voltage stress and cause device failure.In one embodiment, the IGBT di/dt is gradually increased until a spikein the diode voltage reaches the safety limit. This provides the di/dtlevel for the selected operating conditions.

FIG. 6A is a graphical representation 600 of voltage 602 applied to agate with respect to time 604. The graphical representation 600illustrates a first gate drive voltage profile 606 applied to a resistorin series with a gate of an IGBT at a low DC bus voltage (e.g., 200V). Asecond gate drive voltage profile 608 is the applied voltage to aresistor in series with the gate of the IGBT at a high DC bus voltage(e.g., 400V). A first gate-emitter voltage profile 610 applied acrossthe gate and emitter of the IGBT at a low DC bus voltage (e.g., 200V).And, a second gate-emitter voltage profile 612 applied across the gateand emitter of the IGBT at a high DC bus voltage (e.g., 400V). At time614 a gate voltage step function is applied to a first step voltagelevel 616. The gate voltage (606, 608) is maintained at that level untiltime 618, after which the gate voltage (606, 608) is increased at a ratebased on a duration. The duration is derived from IGBT characteristics.After the duration ends at time 618, the gate voltage is increased to anon-voltage. The gate voltage (606, 608) is ramped or increased over atime period that is based on a derivative of the current being less thana threshold corresponding to a supply voltage. For example, At low DCbus voltages the difference is between 618 and 620, and at high DC busvoltages the difference is between 618 and 622.

FIG. 6B is a graphical representation 630 of a current flow to the gate(I_(g)) 632 with respect to time 604. The graphical representation 630illustrates a first gate current profile 634 applied to the gate of theIGBT at the low DC bus voltage and a second gate current profile 636applied to the gate of the IGBT at the high DC bus voltage.

FIG. 6C is a graphical representation 640 of a collector to emittervoltage (V_(ce)) 642 with respect to time 604. The graphicalrepresentation 640 illustrates a first collector to emitter voltageprofile 646 applied to the gate of the IGBT at the low DC bus voltageand a second collector to emitter voltage profile 644 applied to thegate of the IGBT at the high DC bus voltage.

FIG. 6D is a graphical representation 650 of a current flow to thecollector (I_(c)) 652 with respect to time 604. The graphicalrepresentation 650 illustrates a first collector current profile 654flowing to the collector of the IGBT at the low DC bus voltage and asecond collector current profile 656 flowing to the collector of theIGBT at the high DC bus voltage.

The processes, methods, or algorithms disclosed herein can bedeliverable to/implemented by a processing device, controller, orcomputer, which can include any existing programmable electronic controlunit or dedicated electronic control unit. Similarly, the processes,methods, or algorithms can be stored as data and instructions executableby a controller or computer in many forms including, but not limited to,information permanently stored on non-writable storage media such asRead Only Memory (ROM) devices and information alterably stored onwriteable storage media such as floppy disks, magnetic tapes, CompactDiscs (CDs), Random Access Memory (RAM) devices, and other magnetic andoptical media. The processes, methods, or algorithms can also beimplemented in a software executable object. Alternatively, theprocesses, methods, or algorithms can be embodied in whole or in partusing suitable hardware components, such as Application SpecificIntegrated Circuits (ASICs), Field-Programmable Gate Arrays (FPGAs),state machines, controllers or other hardware components or devices, ora combination of hardware, software and firmware components.

While exemplary embodiments are described above, it is not intended thatthese embodiments describe all possible forms encompassed by the claims.The words used in the specification are words of description rather thanlimitation, and it is understood that various changes can be madewithout departing from the spirit and scope of the disclosure. Aspreviously described, the features of various embodiments can becombined to form further embodiments of the invention that may not beexplicitly described or illustrated. While various embodiments couldhave been described as providing advantages or being preferred overother embodiments or prior art implementations with respect to one ormore desired characteristics, those of ordinary skill in the artrecognize that one or more features or characteristics can becompromised to achieve desired overall system attributes, which dependon the specific application and implementation. These attributes mayinclude, but are not limited to cost, strength, durability, life cyclecost, marketability, appearance, packaging, size, serviceability,weight, manufacturability, ease of assembly, etc. As such, embodimentsdescribed as less desirable than other embodiments or prior artimplementations with respect to one or more characteristics are notoutside the scope of the disclosure and can be desirable for particularapplications.

What is claimed is:
 1. A vehicle powertrain comprising: an IGBTconfigured to conduct current between a supply and load; and acontroller configured to apply voltage to a gate of the IGBT at a firstlevel for a first duration that depends on a capacitance of the gate,and to increase the voltage over a second duration based on a rate ofchange of the current falling below a threshold defined by a supplyvoltage for the load.
 2. The vehicle powertrain of claim 1, wherein thefirst level is greater than an IGBT conduction threshold voltage andless than a minimum gate voltage at which the IGBT conducts a maximumload current while the IGBT is in a saturation mode.
 3. The vehiclepowertrain of claim 1, wherein the first duration is directlyproportional to the capacitance of the gate, and inversely proportionalwith a resistance of the gate.
 4. The vehicle powertrain of claim 1,wherein the load is an electric machine or an inductor of a DC-DCconverter.
 5. The vehicle powertrain of claim 4, wherein the secondduration is based on a parasitic inductance and a temperature of theIGBT.
 6. A method of controlling an IGBT of a powertrain invertercomprising: by a gate driver, applying a voltage at a first level onto agate of an IGBT for a predetermined time that is based on a capacitanceof the gate; flowing, by the IGBT in response to the voltage, a currentthrough a collector of the IGBT; and in response to a rate of change ofthe current through the IGBT exceeding a predetermined threshold definedby a supply voltage of the inverter, increasing the voltage from thefirst level to a second level greater than the first level.
 7. Themethod of claim 6, wherein the predetermined time is directlyproportional to the capacitance, and inversely proportional with aresistance of the gate.
 8. The method of claim 6, wherein the firstlevel is greater than a conduction threshold voltage of the IGBT andless than a minimum gate voltage at which the IGBT conducts a maximumload current while the IGBT is in a saturation mode.
 9. The method ofclaim 6, wherein the increasing is over a period of time that is basedon a parasitic inductance and a temperature of the IGBT.
 10. A vehiclecomprising: an IGBT configured to selectively conduct current between asupply and load; and a controller configured to apply a voltage to agate of the IGBT at a first level for a duration derived from aresistance of the gate, and after the duration expires, control a rateof increase of the voltage based on a rate of change of the currentbeing less than a threshold corresponding to a supply voltage.
 11. Thevehicle of claim 10, wherein the first level is further derived from acapacitance of the gate.
 12. The vehicle of claim 11, wherein theduration is directly proportional to the capacitance of the gate, andinversely proportional with a resistance of the gate.
 13. The vehicle ofclaim 10, wherein the rate of increase is based on the current conductedbetween the supply and load.
 14. The vehicle of claim 10, wherein theload is an electric machine or an inductor of a DC-DC converter.